Input circuitry for transistor power amplifier and method for designing such circuitry

ABSTRACT

A circuit having: an input matching network; a transistor coupled to an output of the input matching network; and wherein the input matching network has a first input impedance when such input matching network is fed with an input signal having a relatively low power level and wherein the input matching network has an input impedance different from the first input impedance when such input matching network is fed with an input signal having a relatively high power level.

TECHNICAL FIELD

This invention relates generally to transistor power amplifiers and moreparticularly to input circuitry for such transistor power amplifiers.

BACKGROUND AND SUMMARY

As is known in the art, it is frequently desirable to operate theamplifier linearly and with high efficiency over a wide range of inputpower levels. In order to design such a power amplifier, a fixed inputimpedance network is designed which compromises between performance athigh input power levels and low input power levels.

As is also known in the art, Gallium Nitride (GaN) transistors are highband gap semiconductor devices that operate at high voltages (typically,20 to 50V) and high current densities (up to 1.5 A/mm). The devices havebeen demonstrated to produce 6 W/mm for large periphery devices (over 2mm) over the frequency range from 2 to 20 GHz. At still higherfrequencies up to 40 GHz the transistors have been shown to produce upto 4 W/mm of output power.

One type of GaN includes a Gamma gate, (i.e., a gate with an overhangextending into the gate drain region sometimes also referred to as afield plate), as shown in FIG. 4. The purpose of the overhang is toreduce the fields in that region so that the transistor can be operatedat high voltages to utilize the wide band gap properties of GaN. This isthe type of transistor that is generally used in all our MMIC designapplications at S-, C-, X- and Ku-bands. Transistors such as GaN areused to design high power amplifiers for use in military radars,communication and commercial base station applications. A poweramplifier is usually designed to operate over a specified band offrequencies at a specified power level. At a given frequency within theband, the amplifier response is gauged by its transfer characteristics,i.e., output power versus the input power (drive). As the input drive isincreased the output power is a linear function of the input powerinitially, but eventually reaches saturation or compression.

As is known, small signal means linear operation of a transistor (FET)and large signal means non-linear operation of the transistor. Linearoperation, by definition, means minimal perturbation of a system or anamplifier. For an amplifier, with less than 20 dB of gain that produces1 W of saturated output power, the application of an input drive powerof a few milliwatts would be considered as linear operation. Theresponse of the amplifier is given by a linear equation: Pout=(SSGain)×Pin, where SS Gain is the small signal gain of the amplifier andis a constant in the equation. As Pin is increased this relationshipholds up to an input drive level beyond which SS Gain is replaced by LSGain (large signal gain), which is no longer, a constant. At a certaindrive level LS Gain is lower than SS Gain by 1 dB and is the region inthe Pout vs Pin transfer curve that is called 1 dB compression point.Beyond the 1 dB compression point a normal amplifier will follow thegain response of 1 dB drop in gain for 1 dB increase in Pin (1 dB/1 dB)as shown in FIG. 5. An amplifier that deviates from this gain responsein the large signal region by displaying a slope of 1 dB/2 dB or 1 dB/3dB and so on can be described as having soft compression.

At microwave frequencies where there is spatial and time dependence ofvoltage and current it is customary to use power instead of voltage. Thevoltage is proportional to square root of the power. Each power level ofan amplifier corresponds to a voltage and current. In the small signalregime the voltage and current are sinusoidal. In the large signalregime sinusoidal inputs can lead to an output with distorted voltageand current shapes. So it becomes easier to talk in terms of power. Inthe small signal region the power can be obtained by simplemultiplication of Voltage and Current. In the large signal region Poweris integration of a complex voltage and current over a cycle.

Typical transfer characteristics of a transistor or a power amplifierare shown in FIG. 5, illustrating a linear region at the lower inputpower levels and a hard saturation region at the higher input powerlevels. Also plotted on the same FIG. 5 are solid lines illustratingtransfer characteristics of an ideal transistor. Both curves have acommon X-axis labeled Pin(dBm) (input drive level). The top curve isPout or power out versus Pin. The bottom curve is Gain in dB vs Pin.

GaN based transistors with field plates and power amplifiers operatingat high voltages have been observed to exhibit “soft compression”characteristics depicted by the dotted curve in the same FIG. 5. The 1dB compression of the transistor occurs at Pin=9 dBm, while the “ideal”transistor saturates for Pin=13 dBm or greater. Because of softcompression the non-ideal FET will require a higher input drive level toobtain power saturation of the device and the amplifier designed withsuch FETs will also require a higher drive than normal. Furthermore, a2-stage GaN amplifier will require a conservative choice of the FETratio between stages, thereby contributing to reduced efficiency.

The method used to design an input impedance for the transistor hastypically been as follows: First, the output tuner load is match to 50Ohms. Next, a small signal source pull is used to determine best sourcematch for the best output power. (A source or load pull refers to thetechnique of varying either the input or output match of the transistorsaround the Smith chart until the optimum performance is achieved.)Alternatively, the source match location on the Smith chart can also beobtained from S-parameters of the device at a predetermined frequencywithin the normal operating range of the device, here for example afrequency of 3 GHz. This source match is also called a small signalconjugate match to the input of the device. Next, with this fixed sourcematch, a load pull is performed on the device from low to high drivelevels. The drive level should be high enough to drive the output atleast 3 dB into compression. Next, power and efficiency contours aregenerated from low to high drive levels and the location of the powerand efficiency load targets are noted. Next, Pout versus Pin transfercurves are obtained at the power and efficiency load targets. The systemalso records Gt and Gp (transducer gain and power gain), reflected powerfrom the device input (S11) or return loss among several othermeasurement related parameters. The transfer curves Pout vs Pin clearlyexhibit soft compression characteristics, as shown in FIG. 5. Note theinput drive level required to saturate the device.

Thus, when the input of the GaN devices is matched using theconventional small signal conjugate match or matched at low drive, thedevices exhibit “soft” compression characteristics, rather than the 1dB/1 dB hard compression knee that is desired, as displayed by thetransfer characteristics of Pout vs Pin.

Applicants have discovered that there is significant degradation toamplifier efficiency at high input power levels using a fixed inputimpedance network, (i.e., an input impedance having components which arethe same at both low input power levels and high input power levels).This significant degradation has been determined by the applicants whensuch input impedance networks are coupled to the gate electrode oftransistors having field plates and with GaN transistor poweramplifiers. More particularly, with regard to GaN transistors, suchtransistors have been found to exhibit soft compression (i.e., a gradualtransition between a linear amplification region of the transistor and anon-linear amplification region of the transistor).

Still more particularly, applicants have discovered that softcompression in GaN devices can be significantly reduced or eliminatedusing a matching procedure at the input of the device with relativelylarge input signal drive levels rather than with relatively low signalinput drive levels. More particularly, upon re-matching the device underlarge signal conditions or high drive and then sweeping the transfercurves at a power or efficiency power load, the soft compressionfeatures in the transfer characteristics are significantly removed oreliminated.

In accordance with the present invention, a circuit is provided having:an input matching network; a transistor coupled to an output of theimpedance network; and wherein the input matching network has a firstinput impedance when such input matching network is fed with an inputsignal having a relatively low power level and wherein the inputmatching network has an input impedance different from the first inputimpedance when such input matching network is fed with an input signalhaving a relatively high power level.

In one embodiment, the transistor has a field plate.

In one embodiment, the transistor is a gallium nitride transistor.

In accordance with another feature of the invention, a circuit isprovided having; a transistor having an input electrode; an inputmatching network having an input fed by an input signal and having anoutput connected to the input electrode of the transistors; a powerlevel sensing circuit fed by the input signal; and wherein the inputmatching network is responsive to the power level sensing circuit to:configure the input matching network with a first input impedance whensuch power level sensing circuit senses the input signal has arelatively low power level; and configure the input matching networkwith an input impedance different from the first input impedance whensuch power level sensing circuit senses the input signal has arelatively high power level.

In one embodiment, the input matching network has a first inductorserially coupled between the input signal and the input electrode of thetransistor when such power level sensing circuit senses the input signalhas the relatively high level and wherein the input matching network hasa second inductor serially coupled between the input signal and theinput electrode of the transistor when such power level sensing circuitsenses the input signal has the relatively low power level.

In one embodiment, the input matching network comprises a pair ofelectrical components and at least one switch. The switch operates inresponse to the power level sensing circuit to electrically decouple oneof the pair of electrical components from the input matching network atone of the relatively high or relatively low power levels and operatesto electrically couple said one of the pair of electrical components tothe input matching network at the other one of the relatively high orrelatively low power levels.

The invention thus incorporates an input signal power level dependentelement (i.e., a configurable input matching network). First, an optimalsmall signal input matching network configuration is attached to the GaNtransistor. This provides good stability, return loss, and powertransfer from an RF input to the amplifying transistor at low drivepowers (i.e., low signal power levels), but at the expense of poorperformance under high drive powers (i.e., high signal power levels).The reconfigured input matching network is then used to rotate the phaseangle (i.e., match), only under high input signal power levels, to thatwhat is optimal for realizing peak performance without soft compression.For example, at S-band for a 2.5 mm periphery transistor, this inputmatching network will only have to rotate the original phase angle by 10degree clockwise on the Smith Chart. The reconfigured input matchingnetwork is disconnected from the first via switches in the RF path,activated by a power sensing diode. Under higher drive powers, the powersensing diode and associated circuit would open the RF switches(depletion-mode switch operation), connecting the second matchingnetwork to the first, causing a rotation to the optimal large signalmatch point. The diode's size and bias would be chosen to “turn-on” at aset drive, based on amplifying stage FET periphery. With such anarrangement, the input matching network has a configuration to provideimpedance matching at low input signal drive power levels and adifferent configuration so as to provided impedance matching at the highinput signal drive power levels.

Thus, the invention incorporates a “smart”, tunable or configurableinput matching network with a complex, just now understand, GaN softcompression issue. The invention provides an optimal solution to low andhigh drive stability and performance issues.

In accordance a method is provided for designing an input network for aGaN transistor device. The method includes: driving the device throughthe input network with a relatively large input signal power level;varying parameters of the input network with the output of the device ata predetermined output power level; measuring transfer functionperformance parameters of the device as the input network parameters arevaried; and selecting the input network parameters from the measuredtransfer function performance parameters.

The details of one or more embodiments of the invention are set forth inthe accompanying drawings and the description below. Other features,objects, and advantages of the invention will be apparent from thedescription and drawings, and from the claims.

DESCRIPTION OF DRAWINGS

FIG. 1 is a schematic diagram of a gallium arsenide power amplifieraccording to the invention;

FIG. 2A is an equivalent circuit of the input impedance of acommon-source configured transistor which forms the output stage of theamplifier of FIG. 1 when such amplifier is fed an input signal having arelatively low power level;

FIG. 2B is an equivalent circuit of the input impedance of acommon-source configured transistor which forms the output stage of theamplifier of FIG. 1 when such amplifier is fed an input signal having arelatively high power level;

FIG. 3A is an equivalent circuit of an input matching network coupled toan input of the common-source configured transistor which forms theoutput stage of the amplifier of FIG. 1 when such amplifier is fed aninput signal having a relatively low power level;

FIG. 3B is an equivalent circuit of an input matching network coupled toan input of the common-source configured transistor which forms theoutput stage of the amplifier of FIG. 1 when such amplifier is fed aninput signal having a relatively high power level;

FIG. 4 is a cross-sectional sketch of a GaN transistor device with agamma gate (field plate), such gamma gate being shown enlarged in anencircled region of the transistor;

FIG. 5 are curves of output power (Pout) vs input power (Pin) Transfercharacteristics of the GaN FET of FIG. 1 compared to an “ideal”transistor device;

FIG. 6 is a load pull bench test set-up 10 used to test devices andgenerate the results to be described herein;

FIG. 7 are curves of Pout versus Pin characteristics of a 2.5 mm GaN FETdevice with a conjugate small signal source match (dotted lines) and alarge signal source match (solid lines), the FET being terminated in aload for maximum power;

FIG. 8 are curves of Pout versus Pin characteristics of a 2.5 mm GaN FETdevice with a conjugate small signal source match (dotted lines) and alarge signal source match (solid lines, the FET being terminated in aload for maximum power The device is terminated in a load for maximumefficiency;

FIG. 9 shows the location of the small signal and large signal sourcematches for a device under test on the Smith chart;

FIG. 10 is a small signal model of the GaN transistor device of FIG. 1;and

FIG. 11 is a plot of Cgs versus Vds and Vgs with the large signaldynamic load line superimposed for a device under test.

Like reference symbols in the various drawings indicate like elements.

DETAILED DESCRIPTION

Referring now to FIG. 1, a power amplifier circuit 10 is shown toinclude: an input matching network 12 having an input terminal 13 forcoupling to an RF input signal; an output stage 15 having a transistor14, here a gallium arsenide (GaN) field effect transistor (FET) with afield plate, coupled to an output 16 of the input matching network 12.As will be described in more detail below, the input matching network 12is configurable as a function of the power level, of the input signal.More particularly, the input matching network 12 is configured to have afirst input impedance when such input matching network 12 is fed withthe input signal having a relatively low power level and is configuredto have a different input impedance from the first input impedance whensuch input matching network 12 is an input signal having a relativelyhigh power level.

More particularly, the amplifier circuit 10 includes a power levelsensing circuit 18 connected to input 13 and thus is fed by the inputsignal. As will be described in more detail below, the input matchingnetwork 12 has a first inductor L1 serially coupled between the inputsignal and the input electrode 16 of the transistor 14 when such powerlevel sensing circuit 18 senses the input signal has the relatively highpower level and the input matching network 12 has a second inductor L2serially coupled between the input signal and the input electrode 16 ofthe transistor 14, and hence the inductors L1 and L2 are connected inparallel when such power level sensing circuit 18 senses the inputsignal has the relatively low power level.

The input impedance to the transistor 14 at relatively low input signalpower levels is shown in FIG. 2A and the input impedance to thetransistor 14 at relatively high input signal power levels is shown inFIG. 2B. Thus, at relatively low input signal power levels, the inputimpedance to the transistor 14 is here in this example, a 5.90 ohmresistor in series with a 16.8 pF capacitor (here the transistor 14having a 2.5 mm periphery and series resistor-capacitor (RC) valuesspecified for 3 GHz), while the same transistor 14 operating at the samefrequency but at a relatively high input signal power level, has aninput impedance of a 6.41 ohm resistor in series with a 4.33 pFcapacitor. Thus, the input impedance capacitance changes about 300percent between the high and low power levels of the input signal. Aswill be described in more detail below, at relatively low input signalpower levels, the input matching network 12 is configured as that shownin FIG. 3A and for relatively high input signal power levels, the inputmatching network is configured as that shown in FIG. 3B. It is notedthat when the input matching network 12 is configured for the low inputsignal power level condition, a 0.422 nH inductor, provided by theparallel combination of inductor L1 and L2, is connected in series withthe input impedance of the transistor, as shown in FIG. 3A and when theinput matching network is configured for the high input signal powerlevel condition, a 0.897 nH inductor, provided by inductor L1, isconnected in series with the input impedance of the transistor, as shownin FIG. 3B.

The reconfiguration of the input matching network 12 to either thatshown in FIG. 3A or that shown in FIG. 3B is through the use of thepower level sensing circuit 18 which senses power of the input signaland produces a control signal on line 19 for switches 20, 22, here forexample, depletion-mode FETs Q2 and Q3.

In this schematic, the GaN FET's (i.e., transistor 14) gate impedance,represented as series RC shown in FIG. 2A for low input signal powerlevels and shown in FIG. 2B for high input signal power levels, istransformed through the reconfigurable input matching network 12 to looksimilar to (i.e., impedance matched to) the characteristic systemimpedance; in this case 50 ohms (i.e., the input impedance of theamplifier 10).

More particularly, a coupling network, CN, here for example, shown as acapacitor C8 delivers a portion of the input signal to the power levelsensing circuit 18. It should be understood that the coupling network CNmight be implemented in a number of potential configurations (resistor,capacitor, coupled line . . . etc). This coupling network CN, dependingon the coupling coefficient, has the ability to adjust the threshold forstate change independently of a detector bias network BN, here forexample as resistor R6 connected between a voltage supply (V+) and thejunction 21 between the coupling network CN, here capacitor C8, and adiode D1 of the power level sensing circuit 18. The bias network BNmight be implemented in a number of potential configurations (resistor,inductor, resistor divider . . . etc) also has the ability toindependently adjust threshold for state change, especially through theintroduction of a DC bias voltage, however, the power level sensingcircuit 18 will work quite well passively (no bias). The power levelsensing circuit 18 also includes a capacitor C1. The capacitor C1charges on the negative RF half-cycle through the diode D1, and deliversa low ripple, increasingly negative voltage to a load resistor R5 withan output level related to signal level presented to D1. The voltagepotential across resistor R5 is simultaneously providing a switchingsignal on line 19 to depletion-mode FETs (d-FETs) Q2 and Q3 gates (i.e.,switches 20, 22) through isolation bias resistors R1 and R2. Biasresistors R3 and R4 may be incorporated to ensure proper Q2 and Q3transistors switching. This results in the following behavior: (1) lowRF input signal power levels produces a potential across resistor R5that is below the absolute transistors Q2 and Q3 pinch-off voltage;d-FET drain and source are effectively shorted, and inductors L1 and L2combine in parallel for a reduced value that optimally matches the inputimpedance of transistor 14 and thereby configures the input matchingnetwork to that shown in FIG. 3A; while (2) high levels of RF inputsignal power produces a potential across resistor R5 that is above theabsolute transistors Q2 and Q3 pinch-off voltage, d-FET drain and sourceare effectively opened and inductor L2 is electrically decoupled fromthe input matching network so that inductor L1 alone is seriallyconnected to the input of transistor 14 providing the desired phaseangle rotation and optimally matching the input of transistor 14 throughthe remainder of fixed value components of the input matching network12, here for example, a 50 ohm RF input at 13. Threshold for statechange is controlled by the input signal RF power level, the couplingcoefficient and location of the coupling network CN, the bias networkconfiguration BN and optional bias, the diode D1 output voltagesensitivity, and resistor R5 value. It should be understood that theinductors L1 and L2 could just as easily be capacitors, as a multitudeof input matching networks may be realized utilizing various reactiveand resistive component configurations. In this schematic, thereconfigurable elements are in series with RF signal flow (13 to 16) butcould also be configured in parallel with RF signal flow, again,depending upon input matching network configurations choice—the keytakeaway whether inductive, capacitive or resistive is they are combinedat low level and isolated at high level for a depletion-mode (i.e.,d-FET) switching transistor such as Q2 and Q3. For the parallelreconfigurable element shown, notional inductors and resistors increasein value or notional capacitors reduce in value for a low to high levelRF drive change at the input to the amplifier 10.

The input matching network 12 also includes a non-configurable portion24 having an inductor L5 and a pair of capacitors C6 and C7 arranged asshown.

The output 16 of the input matching network 12 is RF alternating current(AC) coupled to the gate of transistor 14 through a capacitor C2.Biasing to the gate and drain of transistor 14 are provided by thevoltages Vg and Vd, such bias being direct current (DC) coupled to thegate and drain through inductors L3 and L4, respectively, andadditionally AC coupled to ground through capacitors C3 and C4,respectively, as shown to prevent unwanted RF signals at the powersupplies.

Thus amplifier 10 is a 1-stage amplifier with reconfigurable inputmatching network 12 controlled by the power level sensing circuit 18.The gain stage includes DC blocking capacitors C2 and C5 (constrainingthe Vg and Vd bias travel), DC bias injection chokes L3 and L4, as wellas, RF bypass capacitors C3 and C4. The reconfigurable input matchingnetwork 12 includes the partial matching network which includes thefixed value input matching elements, L5, C6 and C7. Switchingtransistors Q2 and Q3, gate isolation resistors R1 and R2, biaspull-down resistors R3 and R4 form a practical switch function. Thepower level sensing circuit 18 includes the RF coupling element (CN), DCbias network (BN), detector circuit containing rectifying diode D1, RFbypass capacitor C1, and load resistor R5. This results in an adjustableoutput signal related to RF input amplitude that triggers Q2 and Q3switching. Switch trigger threshold adjusted by varying CN couplingfactor, BN bias levels, and resistor divider networks utilizingresistors R1/R2 and R5.

Referring now to FIG. 6, a load pull bench set-up 10 used to generatethe results to be described herein is shown. The set-up 10 is acommercial unit manufactured by Maury. The discovery of this inventioncame about during the course of load pull measurements at a frequency of3 GHz on GaN FETs with a field plate. As shown in FIG. 6, the set-upincludes of a device under test (DUT) 12 which in this case the deviceunder test (DUT) 12 was a 2.64 mm (12×220 um) GaN FET device. On eitherside of the device 12 are source and load tuners denoted by 14 and 16.The set-up 10 also includes: an RF source 18, a Reflected Power Sensor20, a Directional Coupler 22, a Gate Bias Tee, 24, a Source Tuner, 26,Input Cables and Probe, 28, a Drain Bias Tee 32, an Output Path, 34 apower Meter 36, a tuner Controller 38 and a Bias System for Gate andDrain 34. The source tuner 14 allows the setting of the source match tothe input of the device 12, while the load tuner 16 can be set to scanvarious output load match conditions. The system software calculates andplots the output load contours both for power and efficiency. Similarly,for fixed output load match the system allows one to determine theoptimum source match.

The laboratory procedure uses the set-up 10 for performing device loadpulls the standard way (or old way) described above, which leads to softcompression, versus a new procedure, to be described, which reduces orremoves soft compression. The Q point of the device is here set to 28Vand 100 ma/mm.

The procedure to design the input impedance for the device 12 is asfollows: As before, the load is match to 50 Ohms, and a small signalsource pull is performed on the device 12 to determine best source matchfor best power. Alternatively, the source match location on the Smithchart can also be obtained from S-parameters of the device at 3 GHz,which is the chosen frequency for this experiment. This source match isalso called a small signal conjugate match to the input of the device12. Next, with this source match fixed, a load pull of the device 12 isperformed from low to high drive levels. The drive level should be highenough to drive the output at least 3 dB into compression. Now, however,unlike the old procedure, the output of the device 12 is at its optimumpower target and a source pull is performed on the device 12 at thelarge signal input drive level and Pout versus Pin transfer curves areobtained at the power and efficiency load targets. The system alsorecords Gt and Gp (transducer gain and power gain), reflected power fromthe device input (S11) or return loss among several other measurementrelated parameters. The new source match (large signal source match)location is found to rotate clockwise about 10 to 15 degrees on theSmith chart from the small signal conjugate match point. The same stepscan be followed to obtain the location of the large signal source matchunder efficiency load conditions. It was found that locations of boththese large signal source match points are within close proximity ofeach other on the Smith chart, so that they can be considered as beingone and the same. Next, this source match is fixed and power sweeps(transfer curves) are performed for both power and efficiency loadtargets. The sweeps indicate that soft compression is considerablyreduced for both load conditions. Transfer curves are obtained forperiodically spaced points between the original small signal sourcematch and the new large signal source match. This same technique hasbeen applied to GaN transistors at X-band, and the same approximately 10degree clockwise rotation of the source input match has been found to betrue.

Thus, a method is provided for designing an input network for a GaNtransistor device. The method includes: driving the device through theinput network E with a relatively large input signal power level;varying parameters of the input network E with the output of the deviceat a predetermined output power level. That is, E is the input tuner,which allows various matches throughout the Smith Chart to be presentedto the device. Likewise H is the output tuner and can also be varied;measuring transfer function performance parameters of the device as theinput network parameters are varied; and selecting the input networkparameters from the measured transfer function performance parameters.

Analysis of Measurements

GaN FETs with field plates have been observed to exhibit soft gaincompression characteristics, the degree of soft compression varying withthe output load impedance presented to the device. For instance, it hasbeen observed that a device matched to a power load impedance has verysoft compression characteristics, while a device matched to anefficiency load exhibits considerably improved compressioncharacteristics. This is illustrated in FIGS. 7 and 8 from Pout versusPin measured load pull data, for power and efficiency load cases. Notethat FIGS. 7 and 8 give measured load pull data of a GaN FET for twodifferent output load match conditions, power and efficiency. Bothfigures plot Pout, Gain and Efficiency versus Pin. The dotted lines ineach figure correspond to small signal input source match conditionsthat result in soft compression. The solid line curves illustrate howthe soft compression is reduced when the input source match is set underlarge signal conditions. Of the two figures, FIG. 7, where the FEToutput is matched to a power load, illustrates the problem of softcompression more clearly. It is less severe in FIG. 8 where the FET ismatched at its output to an efficiency load.

The source match for the dotted curves is a standard small signalconjugate match to the input of the device. The device under measurementwas a 2.5 mm GaN FET with a field/gamma gate biased at 24V, 100 ma/mmand the CW measurement is performed at 3 GHz. Pout versus Pincharacteristics of a 2.5 mm GaN FET with a conjugate small signal sourcematch (dotted lines) and a large signal source match (solid lines). Thedevice is terminated in a load for maximum power.

It is quite apparent, especially from FIG. 7 that under a small signalsource match the device exhibits considerable amount of softcompression. When the input of the device is re-matched under largesignal drive conditions, the compression characteristics are more“normal” as illustrated by the solid lines. A measure of the softcompression is the rate at which the gain drops off with input drive. Ifwe analyze the data in FIG. 7 further, we find that for the standardsource match case (dotted lines) the device hits 1 dB compression pointat Pin=15 dBm. From Pin=15 dBm to 25 dBm the gain falls at the rate of0.3 dB/dB and beyond that at a rate of 0.9 dB/dB. For the large signalsource match condition (solid lines) the device reaches 1 dB compressionat Pin=21 dBm and then the gain drops at the rate of 0.9 dB/dB, which isvery close to the conventionally accepted measure of gain drop-off of 1dB/dB and typically what is observed with GaAs pHEMTs. Note also fromthis data that for the large signal match case the PAE peak occurs withthe device 4.4 dB compressed with a gain of 15.6 dB, while in the smallsignal input match case the gain of the device at the PAE peak is 12.9dB and the device is 7.1 dB comp. This difference has implications inthe way a 2-stage power amplifier (PA) is matched at the input of eachstage, the way the FETs are sized in the amplifier and consequently theimpact on the efficiency of the power amplifier.

The location of the small signal and large signal source matchimpedances is illustrated in the Smith chart in FIG. 9. The sourceimpedance goes through a clockwise rotation of at least 10 degrees fromthe small signal source match location and can be as much as 15 degreesto obtain the best source match under large signal conditions. As thesource match is rotated from small signal to the large signal case for agiven large signal Pin, the degree of soft compression goes from “bad”to normal as illustrated in FIG. 7 earlier. That is the input networkfeeding the transistor is designed by: plotting the complex conjugate ofthe gate to source impedance of the transistor on an impedance Smithchart; and then rotating the plot 10-15 degrees clockwise on theimpedance Smith chart to obtain the input impedance of the inputnetwork.

The soft compression phenomenon is peculiar in general to high voltagedevices (>15V operation) and in particular to GaN devices with a fieldplate. One way to understand this behavior is from the small signalmodel of a GaN FET shown in FIG. 10. The input match of the device iscontrolled to a first order by three intrinsic parametersCgs—gate-source capacitance, Cgd—gate-drain capacitance andgm—transconductance. Rs (source resistance) and Rg (gate resistance)also contribute to the input impedance, but these are consideredparasitics and are not bias dependent. Cgs, Cgd and gm are strongly biasdependent, i.e. they are a function of Vds (drain-source voltage) andVgs (gate-source voltage). For instance, FIG. 11 illustrates how Cgsvaries with Vgs and Vds. When the dynamic loadline for large signaldrive, for class AB operation of the device, is superimposed on the Cgsplane, it is quite apparent that the Cgs value at the Q point and smallsignal match conditions is no longer valid at high drive. In fact we canderive an equivalent large signal Cgs value. A similar analysis wouldhold for Cgd and gm. The performance of the device under large signalconditions could be represented by a new small signal input drivedependent model in which the parameters Cgs, gm and Cgd could berepresented as follows:

Cgs=A1+B1*Pin+C1*Pin² , gm=A2+B2*Pin+C2*Pin² and Cgd=A3+B3*Pin+C3*Pin².

where:

A1 is a constant;

B1 is a constant;

C1 is a constant;

A2 is a constant;

B2 is a constant;

A3 is a constant;

B3 is a constant; and

C3 is a constant.

A number of embodiments of the invention have been described.Nevertheless, it will be understood that various modifications may bemade without departing from the spirit and scope of the invention. Forexample, it should be understood that other arrangements and differentpassive elements of inductors and capacitors may be used for theimpedance matching network so that such network provides impedancematching at both low and high input signal power levels. Further, whilea common-source configuration has been described, the impedance matchingnetwork may be appropriately modified for other transistorconfigurations such as common-gate or common-drain. Further, ifenhancement-mode transistors are used for the switches 20 and 22, theinput matching network would be appropriately modified with otherpassive element configurations to provide impedance matching at both lowand high input signal power levels. Accordingly, other embodiments arewithin the scope of the following claims.

1. A circuit, comprising: an input matching network; a transistorcoupled to an output of the impedance network; and wherein the inputmatching network has a first input impedance when such input matchingnetwork is fed with an input signal having a relatively low power leveland wherein the input matching network has an input impedance differentfrom the first input impedance when such input matching network is fedwith an input signal having a relatively high power level.
 2. Thecircuit recited in claim 1 wherein the transistor has a field plate. 3.The circuit recited in claim 1 wherein the transistor is a galliumnitride transistor.
 4. The circuit recited in claim 3 wherein thetransistor has a field plate.
 5. A circuit, comprising; a transistorhaving an input electrode; an input matching network having an input fedby an input signal and having an output connected to the input electrodeof the transistors; a power level sensing circuit fed by the inputsignal; and wherein the input matching network is responsive to thepower level sensing circuit to: configure the input matching networkwith a first input impedance when such power level sensing circuitsenses the input signal has a relatively low power level; and configurethe input matching network with an input impedance different from thefirst input impedance when such power level sensing circuit senses theinput signal has a relatively high power level.
 6. The circuit recitedin claim 5 wherein the transistor has a field plate.
 7. The circuitrecited in claim 5 wherein the transistor is a gallium nitridetransistor.
 8. The circuit recited in claim 7 wherein the transistor hasa field plate.
 9. The circuit recited in claim 5 wherein the inputmatching network has a first inductor serially coupled between the inputsignal and the input electrode of the transistor when such power levelsensing circuit senses the input signal has the relatively high powerlevel and wherein the input matching network has a second inductorserially coupled between the input signal and the input electrode of thetransistor when such power level sensing circuit senses the input signalhas the relatively low power level.
 10. The circuit recited in claim 9wherein the transistor has a field plate.
 11. The circuit recited inclaim 9 wherein the transistor is a gallium nitride transistor.
 12. Thecircuit recited in claim 11 wherein the transistor has a field plate.13. The circuit recited in claim 9 wherein the transistor is a fieldeffect transistor and wherein the input electrode is a gate electrode ofsuch transistor.
 14. The circuit recited in claim 13 wherein thetransistor has a field plate.
 15. The circuit recited in claim 13wherein the transistor is a gallium nitride transistor.
 16. The circuitrecited in claim 15 wherein the transistor has a field plate.
 17. Thecircuit recited in claim 5 wherein the input matching network includes:a pair of electrical components; and at least one switch, and wherein atleast one switch operates in response to the power level sensing circuitto electrically decouple one of the pair of electrical components fromthe input matching network at one of the relatively high or relativelylow power levels and operates to electrically couple said one of thepair of electrical components to the input matching network at the otherone of the relatively high or relatively low power levels.
 18. Thecircuit recited in claim 17 wherein the electrical components areinductors having different inductances.